Zakładka z wyszukiwarką danych komponentów |
|
AD1890 Arkusz danych(PDF) 11 Page - Analog Devices |
|
AD1890 Arkusz danych(HTML) 11 Page - Analog Devices |
11 / 20 page AD1890/AD1891 REV. 0 –11– Sample Clock Jitter Rejection The loop filter settling time also affects the ability of the AD1890/AD1891 ASRCs to reject sample clock jitter, since the control loop effectively computes a time weighted average or “estimated” new output of many past input and output clock events. This first order low pass filtering of the sample clock ratio provide the AD1890/AD1891 with their jitter rejection characteristic. In the slow settling mode, the AD1890/AD1891 attenuate jitter frequencies higher than 3 Hz ( ≈800 ms for the control loop to settle to an 18-bit “pure” sine wave), and thus reject all but the most severe sample clock jitter; performance is essentially limited only by the FIR filter. In the fast settling mode, the ASRCs attenuate jitter components above 12 Hz ( ≈200 ms for the control loop to settle). Due to the effects of on-chip synchronization of the sample clocks to the 16 MHz (62.5 ns) MCLK master clock, sample clock jitter must be a large percentage of the MCLK period (>10 ns) before perfor- mance degrades in either the slow or fast settling modes. Note that since both past input and past output clocks are used to compute the filtered “current” internal output clock request, jit- ter on both the input sample clock and the output sample clock is rejected equally. In summary: the fast settling mode is best for applications when the sample rates will be dynamically altered (e.g., varispeed situations) while the slow settling mode provides the most sample clock jitter rejection. Clock jitter can be modeled as a frequency modulation process. Figure 7 shows one such model, where a noise source combined with a sine wave source modulates the “carrier” frequency gen- erated by a voltage controlled oscillator. NOISE SOURCE VCO DIGITAL OUT ANALOG IN ADC VOLTAGE SOURCE Σ NOISE WAVEFORM SINE WAVE Figure 7. Clock Jitter Modeled as a Modulated VCO If the jittered output of the VCO is used to clock an analog-to- digital converter, the digital output of the ADC will be contami- nated by the presence of jitter. If the noise source is spectrally flat (i.e., “white” jitter), then an FFT of the ADC digital output would show a spectrum with a uniform noise floor which is el- evated compared to the spectrum with the noise source turned off. If the noise source has distinct frequency components (i.e., “correlated” jitter), then an FFT of the ADC digital output would show symmetrical sidebands around the ADC input sig- nal, at amplitudes and frequencies determined by frequency modulation theory. One notable result is that the level of the noise or the sidebands is proportional to the slope of the input signal, i.e., the worst case occurs at the highest frequency full- scale input (a full-scale 20 kHz sinusoid). The AD1890/AD1891 apply rejection to these jitter frequency components referenced to the input signal. In other words, if a 5 kHz digital sinusoid is applied to the ASRC, depending on the settling mode selected, the ASRC will attenuate sample clock jitter at either 3 Hz above and below 5 kHz (slow settling) or 12 Hz above and below 5 kHz (fast settling). The rolloff is 6 dB per octave. As an example, suppose there was correlated jitter present on the input sample clock with a 1 kHz component, associated with the same 5 kHz sinusoidal input data. This would produce sidebands at 4 kHz and 6 kHz, 3 kHz and 7 kHz, etc., with amplitudes that decrease as they move away from the input signal frequency. For the slow settling mode case, 1 kHz represents more than nine octaves (relative to 3 Hz), so the first two sideband pairs would be attenuated by more than 54 dB. For the fast settling mode case, 1 kHz repre- sents more than seven octaves (relative to 12 Hz), so that the first two sideband pairs would be attenuated by more than 42 dB. The second and higher sideband pairs are attenuated even more because they are spaced further from the input signal frequency. Group Delay Modes The other parameter that determines the likelihood of FIFO in- put overflow or output underflow is the FIFO depth. This is the parameter that is selected by the GPDLYS pin (AD1890 only; this pin is a No Connect for the AD1891). The drawback with increasing the FIFO depth is increasing the device’s overall group delay, but most applications are insensitive to a small in- crease in group delay. [This FIFO-induced group delay is better termed transport delay, since it is frequency independent, and should be kept conceptually distinct from the notion of group delay as used in the polyphase filter bank model. The total group delay of the AD1890/AD1891 equals the FIFO transport delay plus the FIR (polyphase) filter group delay.] In the short group delay mode, the FIFO read and write point- ers are separated by five memory locations ( ≈100 µs equivalent transport delay at a 50 kHz sample rate). This is added to the FIR filter delay (64 taps divided by 2) for a total nominal group delay in short mode of ≈700 µs. The short group delay mode is useful when the input and output sample clocks are asynchro- nous but either do not vary or change very slowly. In the long group delay mode (AD1890 only, the AD1891 is always in the short group delay mode), the FIFO read and write pointers are separated by 96 memory locations ( ≈2 ms equiva- lent transport delay). This is added to the FIR filter delay (64 taps divided by 2) for a total nominal group delay in long mode of ≈3 ms. The long group delay mode is useful when the input and output sample clocks are asynchronous and changing relative to one another, such as during varispeed effects. These delays are deterministic and constant except when FSOUT drops below FSIN which causes the number of FIR filter taps to increase (see “Cutoff Frequency Modification” below). In either mode, if the FIFO read and write addresses cross, the MUTE_O signal will be asserted. Note that in all modes and under all con- ditions, both the highly oversampled low-pass prototype and the polyphase subfilters of the AD1890/AD1891 ASRCs possess a linear phase response. The AD1890 has been designed so that when it is in long group delay mode and fast settling mode, a full 2:1 step change (i.e., occurring between two samples) in sample frequency ratio can be tolerated without output mute. |
Podobny numer części - AD1890 |
|
Podobny opis - AD1890 |
|
|
Link URL |
Polityka prywatności |
ALLDATASHEET.PL |
Czy Alldatasheet okazała się pomocna? [ DONATE ] |
O Alldatasheet | Reklama | Kontakt | Polityka prywatności | Linki | Lista producentów All Rights Reserved©Alldatasheet.com |
Russian : Alldatasheetru.com | Korean : Alldatasheet.co.kr | Spanish : Alldatasheet.es | French : Alldatasheet.fr | Italian : Alldatasheetit.com Portuguese : Alldatasheetpt.com | Polish : Alldatasheet.pl | Vietnamese : Alldatasheet.vn Indian : Alldatasheet.in | Mexican : Alldatasheet.com.mx | British : Alldatasheet.co.uk | New Zealand : Alldatasheet.co.nz |
Family Site : ic2ic.com |
icmetro.com |